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ML13175 ML13176 UHF FM/AM Transmitter
Legacy Device: Motorola MC13175, MC13176
The ML13175 and ML13176 are one chip FM/AM transmitter subsystems designed for AM/FM communication systems. they include a Colpitts crystal reference oscillator, UHF oscillator, / 8 (ML13175) or / 32 (ML13176) prescaler and phase detector forming a versatile PLL system. Targeted applications are in the 260 to 470MHz band and 902 to 982 MHz band covered by FCC Title 47; Part 15. Other applications include local oscillator sources in UHF and 900 MHz receivers, UHF and 900 MHz video transmitters, RF Local Area Networks (LAN), and high frequency clock drivers. ML13175/76 offer the following features; * UHF Current Controlled Oscillator * Uses Easily Available 3rd Overtone or Fundamental Crystals for Reference * Fewer External Parts Required * Low Operating Supply Voltage (1.8 to 5.0 Vdc) * Low Supply Drain Currents * Power Output Adjustable (Up to + 10 dBm ) * Differential Output for Loop Antenna or Balun Transformer Networks * Power Down Feature * ASK Modulated by Switching Output On and Off * (ML13175) fo = 8 x fref, (ML13176) fo = 32 x fref * Operating Temperature Range - TA = -40 to +85C
16 1
ML13175-5P PLASTIC PACKAGE CASE 751B (SO-16)
CROSS REFERENCE/ORDERING INFORMATION PACKAGE LANSDALE MOTOROLA SO 16 SO 16 MC13175D MC13176D ML13175-5P ML13176-5P
Note: Lansdale lead free (Pb) product, as it becomes available, will be identified by a part number prefix change from ML to MLE.
PIN CONNECTIONS
Osc 1 1
Figure 1. Typical Application as 320 MHz AM Transmitter
AM Modulator Osc Tank 1.3k 0.01 Coilcraft 150-05J08 0.165 S2
16 Imod 15 Out Gnd
NC 2 NC 3 Osc 4 4 VEE 5
14 Out 2 13 Out 1 12 VCC 11 Enable 10 Reg. Gnd 9 Xtalb
1 2 3 4
f/N
16 15 14 13
VEE
(1) 150p Z = 50 RFout SMA
ICont 6 PDout 7 Xtale 8
(2) VEE
RFC1
0.1 150p
5 6
1.0k
12 11 10 9
VEE 0.1 27k
VCC
S1
7 8
100p (ML13176) VCC
30p (ML13175)
ML13175-30p ML13176-180p ML13175 Crystal 3rd Overtone 40.0000 MHz
0.01 0.82 1.0k
(3)
ML13176 Crystal Fundamental 10 MHz
VCC
NOTES: 1. 2. 2. 2. 3. 3. 3. 3.
50 coaxial balun, 1/10 wavelength at 320 MHz equals 1.5 inches. Pins 5, 10 & 15 are ground and connected to VEE which is the component/DC ground plane side of PCB. These pins must be decoupled to VCC; decoupling capacitors should be placed as close as possible to the pins. The crystal oscillator circuit may be adjusted for frequency with the variable inductor (ML13175); recommended source is Coilcraft "slot seven" 7mm tuneable inductor , Part #7M3-821. 1.0k resistor. Shunting the crystal prevents it from oscillating in the fundamental mode.
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MAXIMUM RATINGS ( TA = 25 C, unless otherwise noted.)
Rating Power Supply Voltage Operating Supply Voltage Range Junction Temperature Operating Ambient Temperature Storage Temperature Symbol VCC VCC TJ TA Tstg Value 7.0 (max) 1.8 to 5.0 +150 - 40 to + 85 - 65 to +150 Unit Vdc Vdc C C C
ELECTRICAL CHARACTERISTICS (Figure 2; V EE = - 3.0 Vdc, TA = 25 C, unless otherwise noted.)*
Characteristic Supply Current (Power down: I 11 & I16 = 0) Supply Current (Enable [Pin 11] to VCC thru 30 k, I16 = 0) Total Supply Current (Transmit Mode) (Imod = 2.0 mA; f o = 320 MHz) Differential Output Power (f o = 320 MHz; Vref [Pin 9] = 500 mVp-p; f o = N x fref) Imod = 2.0 mA (see Figure 7, 8) Imod = 0 mA Hold-in Range ( fref x N) ML13175 (see Figure 7) ML13176 (see Figure 8) Phase Detector Output Error Current ML13175 ML13176 Oscillator Enable Time (see Figure 22b) Amplitude Modulation Bandwidth (see Figure 24) Spurious Outputs (Imod = 2.0 mA) Spurious Outputs (Imod = 0 mA) Maximum Divider Input Frequency Maximum Output Frequency
* For testing purposes, VCC is ground (see Figure 2).
Pin - - - 13 & 14
Symbol IEE1 IEE2 IEE3 Pout
Min - 0.5 - -
Typ - -14 - 34
Max - -18 -39
Unit A mA mA dBm
2.0 - 13 & 14 f H 3.5 4.0 7 lerror 20 22 11 & 8 16 13 & 14 13 & 14 - 13 & 14 tenable BWAM Pson Psoff fdiv fo - - - - - -
+ 4.7 - 45 6.5 8.0 25 27 4.0 25 - 50 - 50 950 950
- - MHz - - A - - - - - - - - ms MHz dBc MHz
Figure 2. 320 MHz Test Circuit
Imod Osc Tank VEE (1) Coilcraft 150-03J0 8 1 2 3 0.098 4 5 6 27p 10k 7 8 f/N 16 15 0.1 14 13 12 11 10 9 VCC (1) 30k 0.1 Ireg. enable 51 0.01 10k 0.1 RFout 1
51
0.01 RFout 2
0.1
15p (ML13176)
10p (ML13175)
2.2k
ML13175-30p ML13176-33p
0.01 0.82 (3) ML13176 Crystal Fundamental VCC 10 MHz
NOTES: 1. VCC is ground; while V EE is negative with respect to ground. 2. Pins 5, 10 and 15 are brought to the circuit side of the PCB via plated through holes. They are connected together with a trace on the PCB and each Pin is decoupled to VCC (ground). 3. Recommended source is Coilcraft "slot seven inductor " part number 7M3-821.
ML13175 1.0k Crystal 3rd Overtone 40 MHz
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ML13175/ML13176
PIN FUNCTION DESCRIPTIONS
Pin 1&4 Symbol Osc 1, Osc 4 10k 1 0sc 1 Internal Equivalent Circuit VCC 10k 4 Osc 4 Description/External Circuit Requirements CCO Inputs The oscillator is a current controlled type. An external oscillator coil is connected to Pins 1 and 4 which forms a parallel resonance LC tank circuit with the internal capacitance of the IC and with parasitic capacitance of the PC board. Three base-emitter capacitances in series configuration form the capacitance for the parallel tank. These are the base-emitters at Pins 1 and 4 and the base-emitter of the differential amplifier. The equivalent series capacitance in the differential amplifier is varied by the modulating current from the frequency control circuit (see Pin 6, internal circuit). A more thorough discussion is found in the Applications Information section.
5
VEE
VEE 5
Subcon
Supply Ground (VEE) In the PCB layout, the ground pins (also applies to Pins 10 and 15) should be connected directly to chassis ground. Decoupling capacitors to VCC should be placed directly at the ground returns.
VEE 6 ICont VCC
VEE Frequency Control For VCC = 3.0 Vdc, the voltage at Pin 6 is approximately 1.55 Vdc. The oscillator is current controlled by the error current from the phase detector. This current is amplified to drive the current source in the oscillator section which controls the frequency of the oscillator. Figures 9 and 10 show the fosc versus ICont, Figure 5 shows the fosc versus ICont at - 40C, + 25C and + 85C for 320 MHz. The CCO may be FM modulated as shown in Figure 17, ML13176 320 MHz FM Transmitter. A detailed discussion is found in the Applications Information section.
Reg 6 ICont
7
PDout 4.0k
VCC 4.0k
PDout 7
Phase Detector Output The phase detector provides 30 A to keep the CCO locked at the desired carrier frequency. The output impedance of the phase detector is approximately 53 k. Under closed loop conditions there is a DC voltage which is dependent upon the free running oscillator and the reference oscillator frequencies. The circuitry between Pins 7 and 6 should be selected for adequate loop filtering necessary to stabilize and filter the loop response. Low pass filtering between Pin 7 and 6 is needed so that the corner frequency is well below the sum of the divider and the reference oscillator frequencies, but high enough to allow for fast response to keep the loop locked. Refer to the Applications Information section regarding loop filtering and FM modulation.
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PIN FUNCTION DESCRIPTIONS
Pin 8 Symbol Xtale Internal Equivalent Circuit VCC Description/External Circuit Requirements Crystal Oscillator Inputs The internal reference oscillator is configured as a common emitter Colpitts. It may be operated with either a fundamental or overtone crystal depending on the carrier frequency and the internal prescaler. Crystal oscillator circuits and specifications of crystals are discussed in detail in the applications section. With VCC = 3.0 Vdc, the voltage at Pin 8 is approximately 1.8 Vdc and at Pin 9 is approximately 2.3 Vdc. 500 to 1000 mVp-p should be present at Pin 9. The Colpitts is biased at 200 A; additional drive may be acquired by increasing the bias to approximately 500 A. Use 6.2 k from Pin 8 to ground. Regulator Ground An additional ground pin is provided to enhance the stability of the system. Decoupling to the VCC (RF ground) is essential; it should be done at the ground return for Pin 10. 5.0p p 11 Enable 11 Enable Reg Device Enable The potential at Pin 11 is approximately 1.25 Vdc. When Pin 11 is open, the transmitter is disabled in a power down mode and draws less than 1.0 A ICC if the MOD at Pin 16 is also open (i.e., it has no current driving it). To enable the transmitter a current source of 10 A to 90 A is provided. Figures 3 and 4 show the relationship between ICC, VCC and Ireg. enable. Note that ICC is flat at approximately 10 mA for Ireg. enable = 5.0 to 100 A (Imod = 0). Supply Voltage (VCC) The operating supply voltage range is from 1.8 Vdc to 5.0 Vdc. In the PCB layout, the VCC trace must be kept as wide as possible to minimize inductive reactances along the trace; it is best to have it completely fill around the surface mount components and traces on the circuit side of the PCB.
9 9 Xtalb 8
Xtalb
8.0k
12k 4.0k
Xtale
10
Reg. Gnd
VCC
Subcon 8.0k 10 Reg. Gnd 12 VCC VCC 2.4k
12 VCC 13 & 14 Out 1 and Out 2
VCC 15 Out_Gnd Out 1 13 14 Out 2 16 Imod
Differential Output The output is configured differentially to easily drive a loop antenna. By using a transformer or balun, as shown in the application schematic, the device may then drive an unbalanced low impedance load. Figure 6 shows how much the Output Power and Free-Running Oscillator Frequency change with temperature at 3.0 Vdc; Imod = 2.0 mA. Output Ground This additional ground pin provides direct access for the output ground to the circuit board VEE. AM Modulation/Power Output Level The DC voltage at this pin is 0.8 Vdc with the current source active. An external resistor is chosen to provide a source current of 1.0 to 3.0 mA, depending on the desired output power level at a given VCC. Figure 23 shows the relationship of Power Output to Modulation Current, Imod. At VCC = 3.0 Vdc, 3.5 dBm power output can be acquired with about 35 mA ICC. For FM modulation, Pin 16 is used to set the desired output power level as described above. For AM modulation, the modulation signal must ride on a positive DC bias offset which sets a static (modulation off) modulation current. External circuitry for various schemes is further discussed in the Applications Information section.
16
Imod
15 Out_Gnd
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Figure 3. Supply Current versus Supply Voltage
100 10 I CC , SUPPLY CURRENT (mA) 8.0 6.0 4.0 2.0 0 0 1.0 2.0 3.0 VCC, SUPPLY VOLTAGE (Vdc) 4.0 5.0 Ireg. enable = 90 A Imod = 0 I CC , SUPPLY CURRENT (mA)
Figure 4. Supply Current versus Regulator Enable Current
VCC = 3.0 Vdc Imod = 0 10
1.0 0.1
1.0 10 100 Ireg. enable, REGULATOR ENABLE CURRENT (A)
1000
Figure 5. Change Oscillator Frequency versus Oscillator Control Current
f OSC , OSCILLATOR FREQUENCY (MHz) f OSC , OSCILLATOR FREQUENCY (MHz) 10 5.0 0 - 40 C 25 C 85 C 60 - 20 0 20 40 ICont, OSCILLATOR CONTROL CURRENT (A) 80 VCC = 3.0 Vdc Imod = 2.0 mA f = 320 MHz (ICont = 0; TA = 25 C) Free-Running Oscillator 4.0 3.0 2.0 1.0 0 -1.0 - 2.0 - 3.0
Figure 6. Change in Oscillator Frequency and Output Power versus Ambient Temperature
fosc 5.5 PO PO , OUTPUT POWER (dBm) 5.0 4.5 4.0 VCC = 3.0 Vdc Imod = 2.0 mA f = 320 MHz (ICont = 0; TA = 25 C) Free-Running Oscillator 0 50 TA, AMBIENT TEMPERATURE (C) 3.5 3.0 100
- 5.0 -10 -15 - 40
- 4.0 - 50
f ref , REFERENCE OSCILLATOR FREQUENCY (MHz)
41.0 40.8 40.6 40.4 40.2 40.0 39.8 39.6 - 30 - 20 Imod = 2.0 mA ICC = 36 mA PO = 5.4 dBm
Closed Loop Response: VCC = 3.0 Vdc fo = 8.0 x fref Vref = 500 mVp-p
f ref , REFERENCE OSCILLATOR FREQUENCY (MHz)
Figure 7. ML13175 Reference Oscillator Frequency versus Phase Detector Current
Figure 8. ML13176 Reference Oscillator Frequency versus Phase Detector Current
Closed Loop Response: VCC = 3.0 Vdc fo = 32 x fref Vref = 500 mVp-p
10.3
10.2 10.1 10 9.9 9.8 - 30 Imod = 2.0 mA ICC = 35.5 mA PO = 4.7 dBm - 20
Imod = 1.0 mA ICC = 25 mA PO = - 0.2 dBm
Imod = 1.0 mA ICC = 22 mA PO = -1.1 dBm
-10 0 10 20 I7, PHASE DETECTOR CURRENT (A)
30
-10 0 10 20 I7, PHASE DETECTOR CURRENT (A)
30
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Figure 9. Change in Oscillator Frequency versus Oscillator Control Current
f OSC , OSCILLATOR FREQUENCY (MHz) 10 0 -10 - 20 - 30 - 40 -100 VCC = 3.0 Vdc Imod = 2.0 mA TA = 25 C fosc (ICont @ 0) 320 MHz f OSC , OSCILLATOR FREQUENCY (MHz) 20 20 10 0 -10 - 20 - 30 - 40 -100
LANSDALE Semiconductor, Inc.
Figure 10. Change in Oscillator Frequency versus Oscillator Control Current
VCC = 3.0 Vdc Imod = 2.0 mA TA = 25 C fosc (ICont @ 0) 450 MHz
0
400 500 100 200 300 ICont, OSCILLATOR CONTROL CURRENT (A)
600
400 500 0 100 200 300 ICont, OSCILLATOR CONTROL CURRENT (A)
600
Legacy Applications Information APPLICATIONS INFORMATION
EVALUATION PC BOARD The evaluation PCB, shown in Figures 26 and 27, is very versatile and is intended to be used across the entire useful frequency range of this device. The center section of the board provides an area for attaching all SMT components to the component ground side of the PCB (see Figures 28 and 29). Additionally, the peripheral area surrounding the RF core provides pads to add supporting and interface circuitry as a particular application requires. This evaluation board will be discussed and referenced in this section. CURRENT CONTROLLED OSCILLATOR (Pins 1 to 4) It is critical to keep the interconnect leads from the CCO (Pins 1 and 4) to the external inductor symmetrical and equal in length. With a minimum inductor, the maximum free running frequency is greater than 1.0 GHz. Since this inductor will be small, it may be either a microstrip inductor, an air wound inductor or a tuneable RF coil. An air wound inductor may be tuned by spreading the windings, whereas tunable RF coils are tuned by adjusting the position of an aluminum core in a threaded coilform. As the aluminum core coupling to the windings is increased, the inductance is decreased. The temperature coefficient using an aluminum core is better than a ferrite core. The UniCoilTM inductors made by Coilcraft may be obtained with aluminum cores (Part No. 51-129-169). GROUND (Pins 5, 10 and 15) GROUND RETURNS: It is best to take the grounds to a backside ground plane via plated through holes or eyelets at the pins. The application PCB layout implements this technique. Note that the grounds are located at or less than 100 mils from the device pins. DECOUPLING: Decoupling each ground pin to VCC isolates each section of the device by reducing interaction between sections and by localizing circulating currents. LOOP CHARACTERISTICS (Pins 6 and 7) Figure 11 is the component block diagram of the ML1317x PLL system where the loop characteristics are described by the gain constants. Access to individual components of this PLL system is limited, inasmuch as the loop is only pinned out at the phase
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detector output and the frequency control input for the CCO. However, this allows for characterization of the gain constants of these loop components. The gain constants Kp, Ko and Kn are well defined in the ML13175 and ML13176. PHASE DETECTOR (Pin 7) With the loop in lock, the difference frequency output of the phase detector is DC voltage that is a function of the phase difference. The sinusoidal type detector used in the IC has the following transfer characteristic: le = A Sin e The gain factor of the phase detector, Kp (with the loop in lock) is specified as the ratio of DC output current, le to phase error, e: Kp = le/e (Amps/radians) Kp = A Sin e/e Sin e ~ e for e 0.2 radians; thus Kp = A (Amps/radians) Figures 7 and 8 show that the detector DC current is approximately 30 A where the loop loses lock at e = /2 radians; therefore Kp is 30 A/radians. CURRENT CONTROLLED OSCILLATOR, CCO (Pin 6) Figures 9 and 10 show the non-linear change in frequency of the oscillator over an extended range of control current for 320 and 450 MHz applications. Ko ranges from approximately 6.3x105 rad/sec/A or 100 kHz/A (Figure 9) to 8.8x105 rad/sec/A or 140 kHz/A (Figure 10) over a relatively linear response of control current (0 to 100 A). The oscillator gain factor depends on the operating range of the control current (i.e., the slope is not constant). Included in the CCO gain factor is the internal amplifier which can sink and source at least 30A of input current from the phase detector. The internal circuitry at Pin 6 limits the CCO control current to 50 A of source capability while its sink capability exceeds 200 A as shown in Figures 9 and 10. Further information to follow shows how to use the full capabilities of the CCO by addition of an external loop amplifier and filter (see Figure 15). This additional circuitry yields at Ko = 0.145 MHz/A or 9.1x105 rad/sec/A.
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Legacy Applications Information
Figure 11. Block Diagram of ML1317x PLL
i(s) fi = f ref Pins 9,8
Phase Detector Kp = 30 A/rad fn = fo/N
e(s) Pin 7
Low Pass Filter Kf
n(s) = o(s)/N
Pin 6 Amplifier and Current Controlled Oscillator Ko = 0.91Mrad/sec/A fo = nfi Pins 13,14
Divider Kn = 1/N N = 8 : ML13175 N = 32 : ML13176 o(s)
Where: Kp = = Kf = Kn = Ko = = Ko =
Phase detector gain constant in A/rad; Kp = 30 A/rad Filter transfer function 1/N; N = 8 for the MC13175 and 1/N; N = 32 for the MC13176 CCO gain constant in rad/sec/A 9.1 x 105 rad/sec/A
LOOP FILTERING The fundamental loop characteristics, such as capture range, loop bandwidth, lock-up time and transient response are controlled externally by loop filtering. The natural frequency (n) and damping factor ( ) are L important in the transient response to a step input of phase or frequency. For a givenL and lock time wn can be determined from the plot shown in Figure 12.
Figure 12. Type 2 Second Order Response
1.9 1.8 1.7 1.6 1.5 1.4 o (t), NORMALIZED OUTPUT RESPONSE 1.3 1.2 1.1 0.3 0.4 0.5 0.6 0.7 0.8 1.0 1.5 2.0 = 0.1
For L = 0.707 and lock time = 1.0 ms; then = 5.0/t = 5.0 krad/sec. The loop filter may take the form of a simple low pass filter or a lag-lead filter which creates an additional pole at origin in the loop transfer function. This additional pole along with 2 that of the CCO provides two pure integrators (1/s ). In the lag-lead low pass network shown in Figure 13, the values of the low pass filtering parameters R1, R2 and C determine the loop constants n and L. The equations t1=R1 C and t2=R2C are related in the loop filter transfer functions F(s) = 1 + t2s/1 + (t1 +t2)s.
Figure 13. Lag-Lead Low Pass Filter
Vin R1 R2 C VO
0.2
1.0 0.9
0.8
0.7 0.6 0.5 0.4 0.3 0.2 0.1 0 0 1.0 2.0 3.0
The closed loop transfer function takes the form of a 2nd order low pass filter given by, H(s )= KvF(s)/s + KvF(s) From control theory, if the loop filter characteristic has F(0) = 1, the DC gain of the closed loop, Kv is defined as, Kv = KpKoKn and the transfer function has a natural frequency, n = Kv/t1 + t2)1/2 and a dampning factor, L = (n/2) (t2 + 1Kv) Rewriting the above equations and solving for the ML13176 with L = 0.707 and n = 5.0 k rad/sec. Kv = KpKoKn = (30) (0.91 X 106)(1/32) = 0.853 X 106 t1 + t2 = Kv/n2 = 0.853 X 106/(25 X 106) = 34.1 ms t2 = 2L /n = (2)(0.707)/(5 X 103) = 0.283 ms t1 = (Kv/n2) -t2=(34.1-0.283) = 33.8 ms
10 11 12 13
4.0 5.0 6.0 7.0 8.0 9.0 nt
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Legacy Applications Information
For C = 0.47 : then R1 = t1/C = 33.8 X 10 -3/0.47 X 10 -6 = 72 k thus, R2 = t2/C = 0.283 X 10 - 3/0.47 X 10 -6 = 0.60k In the above example, the following standard value components are used, C = 0.47 f; R2 = 620 and R'1 = 72 k - 53 k ~ 18 k (R'1 is defined as R1 - 53 k, the output impedance of the phase detector.) Since the output of the phase detector is high impedance (~50 k) and serves as a current source, and the input to the frequency control, Pin 6 is low impedance (impedance of the two diode to ground is approximately 500 ), it is imperative that the second order low pass filter design above be modified. In order to minimize loading of the R2C shunt network, a higher impedance must be established to Pin 6. A simple solution is achieved by adding a low pass network between the passive second order network and the input to Pin 6. This helps to minimize the loading effects on the second order low pass while further suppressing the sideband spurs of the crystal oscillator. A low pass filter with R3 = 1.0 k and C2 - 1500 p has a corner frequency (fc) of 106 kHz; the reference sideband spurs are down greater than - 60 dBc.
Figure 14. Modified Low Pass Loop Filter
Pin 7 18k R' 1 620 0.47 f R2 C 1.0k R3 C3 VCC 1500pf Pin 6
through 8 are a direct measurement of the hold-in range (i.e. fref x N = fH x 2). Since sin e cannot exceed 1.0, as e approaches /2 the hold-in range is equal to the DC loop gain Kv X N. H = Kv x N where, Kv = KpKoKn. In the above example, H = 27.3 Mrad/sec fH = 4.35 MHz EXTENDED HOLD-IN RANGE The hold-in range of about 3.4% could cause problems over temperature in cases where the free-running oscillator drifts more than 2 to 3% because of relatively high temperature coefficients of the ferrite tuned CCO inductor. This problem might worsen for lower frequency applications where the external tuning coil is large compared to internal capacitance at Pins 1 and 4. To improve hold-in range performance, it is apparent that the gain factors involved must be carefully considered. Kn = is either 1/8 in the ML13175 or 1/32 in the ML13176 Kp = is fixed internally and cannot be altered. Ko = Figures 9 and 10 suggest that there is capability of greater control range with more current swing. However, this swing must be symmetrical about the center of the dynamic response. The suggested zero current operating point for 100 A swing of the CCO is at about + 70 A offset point. Ka = External loop amplification will be necessary since the phase detector only supplies 30 A. In the design example in Figure 15, an external resistor (R5) of 15 k to VCC (3.0 Vdc) provides approximately 100 A of current boost to supplement the existing 50 A internal source current. R4 (1.0 k) is selected for approximately 0.1 Vdc across it with 100 A. R1, R2 and R3 are selected to set the potential at Pin 7 and the base of 2N4402 at approximately 0.9 Vdc and the emitter at 1.55 Vdc when error current to Pin 6 is approximately zero A. C1 is chosen to reduce the level of the crystal sidebands.
HOLD-IN RANGE The hold-in range, also called the lock range, tracking range and synchronization range, is the ability of the CCO frequency, fo to track the input reference signal, fref * N as it gradually shifted away from the free running frequency, ff. Assuming that the CCO is capable of sufficient frequency deviation and that the internal loop amplifier and filter are not overdriven, the CCO will track until the phase error, e approaches /2 radians. Figures 5
Figure 15. External Loop Amplifier
VCC = 3.0Vdc 12 C1 1000p R1 7 30A R2 33k 5, 10, 15 68k R3 4.7k R4 1.0k 2N4402 R5 15k 1.6V 6 50A Oscillator Control Circuitry
30A Phase Detector Output
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Legacy Applications Information
Figure 16 Shows the improved hold-in range of the loop. The fref is moved 950 kHz with over 200 A swing of control current for an improved hold-in range of 15.2 MHz or 95.46 Mrad/sec.
f ref , REFERENCE OSCILLATOR FREQUENCY (MHz)
fc = 0.159/RC; For R = 1.0 k + R7 (R7 = 53 k) and C = 390 pF fc = 7.55 kHz or c = 47 krad/sec The application example in Figure 17a of a 320 MHz FM transmitter demonstrates the FM capabilities of the IC. A high value series resistor (100 k) to Pin 6 sets up the current source to drive the modulation section of the chip. Its value is dependent on the peak to peak level of the encoding data and the maximum desired frequency deviation. The data input is AC coupled with a large coupling capacitor which is selected for the modulating frequency. The component placements on the circuit side and ground side of the PC board are shown in Figures 28 and 29 respectively. For voice application using a dynamic or an electret microphone, an op amp is used to amplify the microphone's low level output . The microphone amplifier circuit is shown in Figure 19. Figure 17b shows an application example for NBFM audio or direct FSK in which the reference crystal oscillator is modulated.
Figure 19. Microphone Amplifier
Data Input VCC 3.3k Voice Input Electret Microphone 1.0k 1.0 MC33171
Figure 16. ML13176 Reference Oscillator Frequency versus Oscillator Control Current
Closed Loop Response: fo = 32 x fref VCC = 3.0 Vdc ICC = 38 mA Pout = 4.8 dB Imod = 2.0 mA Vref = 500 mVp-p
10.6 10.4 10.2 10 9.8 9.6 9.4 -150
-100 - 50 0 50 I6, OSCILLATOR CONTROL CURRENT (A)
100
LOCK-IN RANGE/CAPTURE RANGE If a signal is applied to the loop not equal to free running frequency, ff, then the loop will capture or lock-in the signal by making fs = fo (i.e. if the initial frequency difference is not too great). The lock-in range can be expressed as L ~ 2L n FM MODULATION Noise external to the loop (phase detector input) is minimized by narrowing the bandwidth. This noise if minimal in a PLL system since the reference frequency is usually derived from a crystal oscillator. FM can be achieved by applying a modulation current superimposed on the control current of the CCO. The loop bandwidth must be narrow enough to prevent the loop from responding to the modulation frequency components, thus, allowing the CCO to deviate in frequency. The loop bandwidth is related to the natural frequency wn. In the lag-lead design example where the natural frequency, n = 5.0 krad/sec and a damping factor, L = 0.707, the loop bandwidth = 1.64 kHz. Characterization data of the closed loop responses for both the ML13175 and ML13176 at 320 MHz (Figures 7 and 8, respectively) show satisfactory performance using only a simple low-pass loop filter network. The loop filter response is strongly influenced by the high output impedance of the push-pull current output of the phase detector.
100k 120k VCC 3.9k 10k 10k
Data or Audio Output
LOCAL OSCILLATOR APPLICATION To reduce internal loop noise, a relatively wide loop bandwidth is needed so that the loop tracks out or cancels the noise. This is emphasized to reduce inherent CCO and divider noise or noise produced by mechanical shock and environmental vibrations. In a local oscillator application the CCO and divider noise should be reduced by proper selection of the natural frequency of the loop. Additional low pass filtering of the output will likely be necessary to reduce the crystal sideband spurs to a minimal level.
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Legacy Applications Information
Figure 17a. 320 MHz ML13176 FM Transmitter
RF Level Adjust
Osc Tank 1.1k
1 2
16 0.047 15 14 13 RFC1 (3) 12 11 (1)
5.0k CW
VCC
Coilcraft 146-04J08
3 0.146
VCC = 3.8 to 3.3 Vdc
SMA
50 4 510p VCC VCC f/32
RF Output to Antenna
0.1 9.1k 15k 1.0k 18k 2N4402 100k 8 7
(2) VEE
0.47
5 6
130k
620
27k 10 0.47 9 VCC
33k
Data Input (1.6 Vp-p)
51p
51p
220p 6.8 (4)
Crystal Fundamental 10 MHz
(5)
NOTES: 1. 50 coaxial balun, 2 inches long. 2. Pins 5, 10 and 15 are grounds and connnected to VEE which is the component's side ground plane. These pins must be decoupled to VCC; decoupling capacitors should be placed as close as possible to the pins. 3. RFC1 is 180 nH Coilcraft surface mount inductor or 190 nH Coilcraft 146-05J08. 4. Recommended source is a Coilcraft TMslot seven 7.0 mm tuneable inductor part #7M3-682. , 5. The crystal is a parallel resonant, fundamental mode calibrated with 32 pF load capacitance.
Figure 17b. 320 MHz NBFM Transmitter
RF Level Adjust
Osc Tank 1.0k
1 2
16 0.047 15 14 13 RFC1 (3) 12 11 (1)
5.0k CW
VCC
Coilcraft 146-04J08
3 0.146 4 VCC 4700p 9.1k 15k 1.0k 15k 2N4402 33k 8
External Loop Amp
SMA
UT-034 470p f/32
RF Output to Antenna
0.1
(2) VEE
5 6
VCC (3.6 Vdc - Lithium Battery) VCC
130k
6.2k
27k 7 10 9 VCC 10p 100p 180p (6)
Crystal Fundamental 10MHz
0.47 RFC2 (4) RFC3 VCC 1.0k 10 + 0.01
(5) MMBV432L
NOTES: 1. 50 coaxial balun, 2 inches long. 2. Pins 5, 10 and 15 are grounds and connnected to VEE which is the component's side ground plane. These pins must be decoupled to VCC; decoupling capacitors should be placed as close as possible to the pins. 3. RFC1 is 180 nH Coilcraft surface mount inductor. 4. RFC2 and RFC3 are high impedance crystal frequency of 10 MHz; 8.2 H molded inductor gives XL > 1000 .. 5. A single varactor like the MV2105 may be used whereby RFC 2 is not needed. 6. The crystal is a parallel resonant, fundamental mode calibrated with 32 pF load capacitance.
Audio or Data Input
Page 10 of 16
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LANSDALE Semiconductor, Inc.
ML13175/ML13176
Legacy Applications Information REFERENCE CRYSTAL OSCILLATOR (Pins 8 and 9) Selection of Proper Crystal: A crystal can operate in a number of mechanical modes. The lowest resonant frequency mode is its fundamental while higher order modes are called overtones. At each mechanical resonance, a crystal behaves like a RLC series-tuned circuit having a large inductor and a high Q. The inductor Ls is series resonance with a dynamic capacitor, Cs determined by the elasticity of the crystal lattice and a series resistance Rs, which accounts for the power dissipated in heating the crystal. This series RLC circuit is in parallel with a static capacitance, Cp which is created by the crystal block and by the metal plates and leads that make contact with it. Figure 20 is the equivalent circuit for a crystal in a signal resonant mode. It is assumed that other modes of resonance are so far off frequency that their effects are negligible. Series resonant frequency, fs is given by; fs = 1/2(LsCs)1/2 and parallel resonant frequency, fp is given by; fp = fs(1 + Cs/Cp)1/2
Figure 20. Crystal Equivalent Circuit
L3 Cp R3 C3
the frequency separation at resonance is given by; f = fp-fs = fs[1 - (1+ Cs/Cp)1/2] Usually fp is less than 1% higher than fs, and a crystal exhibits an extremely wide variation of the reactance with frequency between fp and fs. A crystal oscillator circuit is very stable with frequency. This high rate of change of impedance with frequency stabilizes the oscillator, because any significant change in oscillator frequency will cause a large phase shift in the feedback loop keeping the oscillator on frequency.
Page 11 of 16
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ML13175/ML13176
LANSDALE Semiconductor, Inc.
Legacy Applications Information
Manufacturers specify crystal for either series or parallel resonant operation. The frequency for the parallel mode is calibrated with a specified shunt capacitance called a "load capacitance." The most common value is 30 to 32 pF. If the load capacitance is placed in series with the crystal, the equivalent circuit will be series resonance at the specified parallel-resonant frequency. Frequencies up to 20 MHz use parallel resonant crystal operating in the fundamental mode, while above 20 MHz to about 60 MHz, a series resonant crystal specified and calibrated for operation in the overtone mode is used. APPLICATION EXAMPLES Two types of crystal oscillator circuits are used in the applications circuits: 1) fundamental mode common emitter Colpitts (Figures 1, 17a, 17b and 21) and 2) third overtone impedance inversion Colpitts (also Figures 1 and 21). The fundamental mode common emitter Colpitts uses a parallel resonant crystal calibrated with a 32 pf load capacitance. The capacitance values are chosen to provide excellent frequency stability and output power of > 500 mVp-p at Pin 9. In Figures 1 and 21, the fundamental mode reference oscillator is fixed tuned relying on the repeatability of the crystal and passive network to maintain the frequency, while in the circuit shown in Figure 17, the oscillator frequency can be adjusted with the variable inductor for the precise operating frequency. The third overtone impedance inversion Colpitts uses a series resonance crystal with a 25 ppm tolerance. In the application examples (Figures 1 and 21), the reference oscillator operates with the third overtone crystal at 40.0000 MHz. Thus, the ML13175 is operated at 320 MHz (fo/8 = crystal; 320/8) = 40.0000 MHz. The resistor across the crystal ensures that the crystal will operate in the series resonance mode. A tuneable inductor is used to adjust the oscillation frequency; it forms a parallel resonant circuit with the series and parallel combination of the external capacitors forming the divider and feedback network and the base-emitter capacitance of the devices. If the crystal is shorted, the reference oscillator should free-run at the frequency dictated by the parallel resonant LC network. The reference oscillator can be operated as high as 60 MHz with a third overtone crystal. Therefore, it is possible to use the ML13175 up to at least 480 MHz and the ML13176 up to 950 MHz (based on the maximum capability of the divider netowork). ENABLER (Pin 11) The enabling resistor at Pin 11 is calculated by: Reg. enable = VCC - 1.0 Vdc/lreg. enable From Figure 4, lreg.enable is chosen to be 75A. So, for a VCC = 3.0 Vdc Rreg.enable = 26.6 k, a standard value 27 k resistor is adequate. LAYOUT CONSIDERATIONS Supply (Pin 12): In the PCB layout the VCC trace must be kept as wide as possible to minimize inductive reactance along the trace; it is best that VCC (RF ground) completely fills around the surface mounted components and interconnect traces on the circuit side of the board. This technique is demonstrated in the evaluation PC board. BATTERY/SELECTION/LITHIUM TYPES The device may be operated from a 3.0 V lithium battery. Selection of a suitable battery is important. Because one of the major problems for long life battery powered equipment is oxidation of the battery terminals, a battery mounted in clip-in socket is not advised. The battery leads or contact post should be isolated from the air to eliminate oxide build-up. The battery should have PC board mounting tabs which can be soldered to the PCB. Consideration should be given for the peak current capability of the battery. Lithium batteries have current handling capabilities based on the composition of the lithium compound, construction and the battery size. A 1300 mA/hr rating can be achieved in the cylindrical cell battery. The Rayovac CR2/3A lithium-manganese dioxide battery is a crimp sealed, spiral wound 3.0 Vdc, 1300 mA/hr cylindrical cell with PC board mounting tabs. It is an excellent choice based on capacity and size (1.358" long by 0.665" in diameter). DIFFERENTIAL OUTPUT (Pins 13, 14) The availability of micro-coaxial cable and small baluns in surface mount and radial-leaded components allows for simple interface to the output ports. A loop antenna may be directly connected with bias via RFC or 50 resistors. Antenna configuration will vary depending on the space available and the frequency of operation. AM MODULATION (Pin 16) Amplitude Shift Key: The ML13175 and ML13176 are designed to accommodate Amplitude Shift Keying (ASK). ASK modulation is a form of digital modulation corresponding to AM. The amplitude of the carrier is switched between two or more values in response to the PCM code. For the binary case, the usual choice is On-Off Keying (often abbreviated OOK). The resultant amplitude modulated waveform consists of RF pulses called marks, representing binary 1 and spaces representing binary 0.
Page 12 of 16
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LANSDALE Semiconductor, Inc.
ML13175/ML13176
Legacy Applications Information
Figure 21. ASK 320 MHz Application Circuit
Rmod 3.3k 1 2 3 0.165 4 (2) VEE 150p 5 6 1.0k 7 8 10 9 f/N 13 RFC1 12 11 VEE VCC 27k S1 0.1 (5) 16 0.01 15 14 VEE (1) Z = 50 150p SM A RFOut
Osc Tank Coilcraft 150-05J08
(4) On-Off Keyed Input TTL Level 10 kHz
0.1
100p (ML13176)
VCC
30p (ML13175)
ML13175-30p ML13176-180p 0.82 ML13175 Crystal 1.0k 3rd Overtone 40.0000 MHz
0.01 (3) ML13176 Crystal Fundamental 10 MHz VCC
NOTES: 1. 50 coaxial balun, 1/10 wavelength line (1.5" ) provides the best match to a 50 load. 2. Pins 5, 10 and 15 are ground and connnected to VEE which is the component/DC ground plane side of PCB. These pins must be decoupled to VCC; decoupling capacitors should be placed as close as possible to the pins. 3. The crystal oscillator circuit may be adjusted for frequency with the variable inductor (MC13175); 1.0 k resistor shunting the crystal prevents it from oscillating in the fundamental mode. Recommended source is Coilcraft "slot seven" 7.0 mm tuneable inductor, part #7M3-821.
4. The On-Off keyed signal turns the output of the transmitter off and on with TTL level pulses through R mod at Pin 16. The "On" power and I CC is set by the resistor which sets Imod = VTTL - 0.8 / Rmod. (see Figure 23). 5. S1 simulates an enable gate pulse from a microprocessor which will enable the transmitter. (see Figure 4 to determine precise value of the enabling resistor based on the potential of the gate pulse and the desired enable.)
Figure 21 shows a typical application in which the output power has been reduced for linearity and current drain. The current draw on the device is 16 mA ICC (average) and -22.5 dBm (average power output) using a 10 kHz modulating rate for the on-off keying. This equates to 20 mA and -2.3 dBm "on", 13 mA and -41 dBm "Off ". The crystal oscillator
enable time is needed to set the acquisition timing. It takes typically 4.0 msec to reach full magnitude of the oscillator waveform. A square waveform of 3.0 V peak with a period that is greater than the oscillator enable time is applied to the Enable (Pin 11)
Page 13 of 16
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ML13175/ML13176
LANSDALE Semiconductor, Inc.
Legacy Applications Information
Figure 23. Power Output versus Modulation Current
10 5.0 PO, POWER OUTPUT (dBm) 0 VCC = 3.0 Vdc f = 320 MHz
- 5.0 -10 -15
- 20 - 25 0.1 1.0 Imod, MODULATION CURRENT (mA) 10
ANALOG AM In analog AM applications, the output amplifier's linearity must be carefully considered. Figure 23 is a plot of Power Output versus Modulation Current at 320 MHz, 3.0 Vdc. In order to achieve a linear encoding of the modulating sinusoidal waveform on the carrier, the modulating signal must amplitude modulate the carrier in the linear portion of its power output response. When using a sinewave modulating signal, the signal rides on a positive DC offset called Vmod which sets a static (modulation off) modulation current, Imod. Imod controls the power output of the IC. As the modulating signal moves around this static bias point the modulating current varies causing power output to vary or to be AM modulated. When the IC is operated at modulation current levels greater than 2.0 mAdc the differential output stage starts to saturate.
Page 14 of 16
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Issue c
ML13175/ML13176
LANSDALE Semiconductor, Inc.
Legacy Applications Information
In the design example, shown in FIgure 24, the operating point is selected as a tradeoff between average power output and quality of the AM. For VCC = 3.0 Vdc;ICC = 18.5 mA and Imod = 0.5 mAdc and a static DC offset of 1.04 Vdc, the circuit shown in Figure 24 completes the design. Where Rmod = (VCC - 1.04 Vdc)/0.5 mA = 3.92 k, use a standard value resistor of 3.9 k.
Figure 24. Analog AM Transmitter
3.9k 1.04Vdc 560 VCC 16 3.0Vdc Rmod 0.8Vdc Data Input 800mVp-p + 6.8
Page 15 of 16
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Issue c
ML13175/ML13176
LANSDALE Semiconductor, Inc.
OUTLINE DIMENSIONS
PLASTIC PACKAGE (ML13175-5P, ML13176-5P) CASE 751B (SO-16)
NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: MILLIMETER. 3. DIMENSION A AND B DO NOT INCLUDE MOLD PROTRUSION. 4. MAXIMUM MOLD PROTRUSION 0.15 (0.006) PER SIDE. 5. 751B-01 IS OBSOLETE, NEW STANDARD 751B-03. DIM A B C D F G J K M P R MILLIMETERS MIN MAX 9.80 10.00 4.00 3.80 1.75 1.35 0.49 0.35 1.25 0.40 1.27 BSC 0.25 0.19 0.25 0.10 5.80 0.25 6.20 0.50 INCHES MIN MAX 0.386 0.393 0.150 0.157 0.054 0.068 0.014 0.019 0.016 0.049 0.050 BSC 0.008 0.009 0.004 0.009 0.229 0.010 0.244 0.019
-A -
16 9
1
-B -
8
P
8 PL
0.25 (0.010)
M
B
M
R X 45 G -T - D 16 PL 0.25 (0.010)
M
C
SEATING PLANE
K TB
S
M
F
J
A
S
Lansdale Semiconductor reserves the right to make changes without further notice to any products herein to improve reliability, function or design. Lansdale does not assume any liability arising out of the application or use of any product or circuit described herein; neither does it convey any license under its patent rights nor the rights of others. "Typical" parameters which may be provided in Lansdale data sheets and/or specifications can vary in different applications, and actual performance may vary over time. All operating parameters, including "Typicals" must be validated for each customer application by the customer's technical experts. Lansdale Semiconductor is a registered trademark of Lansdale Semiconductor, Inc.
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Issue c

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